Transmitter and method of transmitting

ABSTRACT

A transmitter transmitting payload data using OFDM symbols includes a frame builder configured to receive the payload data and to receive signalling data for use in detecting and recovering the payload data at a receiver, and to form the payload data and the signalling data into frames for transmission. A modulator can modulate a first OFDM symbol with the signalling data forming a first of the frames and modulate one or more second OFDM symbols with the payload data forming one or more other frames, and a transmission unit transmits the first and second OFDM symbols. The first OFDM symbol is combined before transmission with a signature sequence that can be configured to allow for detection of the first OFDM symbol at the receiver and decoding the signalling data before the one or more second OFDM symbols carrying the payload data and at lower signal to noise ratios.

FIELD OF THE DISCLOSURE

The present disclosure relates to transmitters and methods oftransmitting payload data using Orthogonal Frequency DivisionMultiplexed (OFDM) symbols.

BACKGROUND OF THE INVENTION

There are many examples of radio communication systems in which data iscommunicated using Orthogonal Frequency Division Multiplexing (OFDM).Systems which have been arranged to operate in accordance with DigitalVideo Broadcasting (DVB) standards for example, use OFDM. OFDM can begenerally described as providing K narrow band sub-carriers (where K isan integer) which are modulated in parallel, each sub-carriercommunicating a modulated data symbol such as Quadrature AmplitudeModulated (QAM) symbol or Quadrature Phase-shift Keying (QPSK) symbol.The modulation of the sub-carriers is formed in the frequency domain andtransformed into the time domain for transmission. Since the datasymbols are communicated in parallel on the sub-carriers, the samemodulated symbols may be communicated on each sub-carrier for anextended period, which can be longer than the coherence time of theradio channel. The sub-carriers are modulated in parallelcontemporaneously, so that in combination the modulated carriers form anOFDM symbol. The OFDM symbol therefore comprises a plurality ofsub-carriers each of which has been modulated contemporaneously withdifferent modulation symbols. During transmission, a guard intervalfilled by a cyclic prefix of the OFDM symbol precedes each OFDM symbol.When present, the guard interval is dimensioned to absorb any echoes ofthe transmitted signal that may arise from multipath propagation orother transmitters transmitting the same signal from a differentgeographic location.

As indicated above, the number of narrowband carriers K in an OFDMsymbol can be varied depending on operational requirements of acommunications system. The guard interval represents overhead and so maybe minimized as a fraction of the OFDM symbol duration in order toincrease spectral efficiency. For a given guard interval fraction, theability to cope with increased multipath propagation whilst maintaininga given spectral efficiency can be improved by increasing the number Kof sub-carriers thereby increasing the duration of the OFDM symbol.However, there can also be a reduction in robustness in the sense thatit may be more difficult for a receiver to recover data transmittedusing a high number of sub-carriers compared to a smaller number ofsub-carriers, because for a fixed transmission bandwidth, increasing thenumber of sub-carriers K also means reducing the bandwidth of eachsub-carrier. A reduction in the separation between sub-carriers can makedemodulation of the data from the sub-carriers more difficult forexample, in the presence of Doppler frequency shifts. That is to saythat although a larger number of sub-carriers (high order operatingmode) can provide a greater spectral efficiency, for some propagationconditions, a target bit error rate of communicated data may require ahigher signal to noise ratio to achieve than required for a lower numberof sub-carriers.

SUMMARY OF DISCLOSURE

According to an aspect of the present disclosure there is provided atransmitter for transmitting payload data using Orthogonal FrequencyDivision Multiplexed (OFDM) symbols. The transmitter comprises a framebuilder configured to receive the payload data to be transmitted and toreceive signalling data for use in detecting and recovering the payloaddata at a receiver, and to form the payload data and with the signallingdata into frames for transmission. Each frame includes in one part thesignalling data and in another part the payload data. A modulator isconfigured to modulate a first OFDM symbol with the signalling data foreach frame and to modulate one or more second OFDM symbols with thepayload data, a combiner to combine the first OFDM symbol with asignature sequence and a transmission unit transmits the first andsecond OFDM symbols. Accordingly embodiments of the present techniqueare arranged so that the first OFDM symbol which carries the signallingdata is combined before transmission with the signature sequence. Thesignature sequence is configured to allow detection of the first symbolof the frame at the receiver and decoding of the signalling data atlower signal to noise ratios than may be required for the payload data.

Embodiments of the present disclosure can provide a transmitter, whichis arranged to transmit payload data using Orthogonal Frequency DivisionMultiplexing (OFDM) symbols. The transmitter comprises a frame builderwhich is adapted to receive the payload data to be transmitted and toreceive signalling data for use in detecting and recovering the data tobe transmitted at a receiver. The frame builder is configured to formthe payload data and the signalling data into frames for transmission. Amodulator is configured to modulate the payload data and the signallingdata onto OFDM symbols, and a transmission unit is arranged to transmitthe OFDM symbols. The signalling data is formed into the frame andtransmitted using a first OFDM symbol and the payload data is formedinto one or more other frames and transmitted using one or more secondtype of OFDM symbol in accordance with transmission parameters, such asa coding rate, a modulation scheme and an operating mode for the numberof sub-carriers for OFDM symbols. The transmission parameters for thesecond type of OFDM symbol may be included within the signalling data.Thus the signalling data may be detected first by a receiver in order torecover the payload data. In order to facilitate detection of the firstOFDM symbol, carrying the signalling data in challenging receptionenvironments the first OFDM symbol is combined before transmission witha signature sequence for that can be used by the receiver to identifythe first OFDM symbol within the frame.

Embodiments of the present disclosure can provide an arrangement inwhich a signature sequence is combined with OFDM symbols carrying, forexample, signalling data so that there is an improved likelihood of areceiver being able to detect the OFDM symbols carrying the signallingdata. The OFDM symbols carrying signalling data will be referred to assignalling OFDM symbols and in one example may form a preamble part of atransmission frame in which payload data is transmitted using other OFDMsymbols.

According to an arrangement in which embodiments of the presentdisclosure find application there is a requirement to provide a“preamble” OFDM symbol in a transmission frame, which carries signallingparameters to indicate, for example, at least some of the communicationsparameters which were used to encode and to modulate payload data ontothe data bearing OFDM symbols whereby after detecting the signallingdata within the first (preamble) OFDM symbol the receiver can recoverthe transmission parameters in order to detect the payload data from thedata bearing OFDM symbols.

In the following description the first OFDM symbol may be a preambleOFDM symbol or form part of one in a transmission frame and so may bereferred to as a preamble OFDM symbol and because this is arranged tocarry signalling data, it may be referred to as a signalling OFDMsymbol.

According to one embodiment a number of sub-carriers used for the OFDMsymbols carrying signalling data may be different from the number ofsub-carriers used for the OFDM symbols which are used to carry thepayload data. For example in order to improve a likelihood of recoveringthe signalling data, making it more robust for detection in morechallenging radio environments the number of sub-carriers may be smallerthan for the OFDM symbols carrying payload data. For example, thepayload data bearing OFDM symbols may be required to have a highspectral efficiency and therefore for example the number of sub-carriersmay be 16 k (16384) or 32 k (32768) whereas in order to improve alikelihood that a receiver can recover the signalling data from thesignalling OFDM symbols, the number of sub-carriers for the firstsignalling OFDM symbol may be a lower number, for example, a 4 k (4096)or 8 k (8192).

Various further aspects and features of the disclosure are defined inthe appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present disclosure will now be described by way ofexample only with reference to the accompanying drawings wherein likeparts are provided with corresponding reference numerals and in which:

FIG. 1 is a schematic diagram illustrating an arrangement of a broadcasttransmission network;

FIG. 2 is a schematic block diagram illustrating an example transmissionchain for transmitting broadcast data via the transmission network ofFIG. 1;

FIG. 3 is a schematic illustration of OFDM symbols in the time domainwhich include a guard interval;

FIG. 4 is a schematic block of a typical receiver for receiving databroadcast by the broadcast transmission network of FIG. 1 using OFDM;

FIG. 5 is a schematic illustration of a transmission frame fortransmitting broadcast data including payload data and signalling data;

FIG. 6 is a block diagram showing a transmitter for transmittingsignalling data via a signalling or preamble OFDM symbol according toone embodiment;

FIG. 7 is a schematic block diagram of a signature sequence generatoraccording to one embodiment;

FIG. 8 is a graphical plot of bit error rate with respect to signal tonoise ratio in the presence of additive white Gaussian noise for codingrates of one half and one quarter;

FIG. 9 is a graphical plot of bit error rate with respect to a signaturesequence back-off power from the power of the modulated signalling datawhich provides an acceptable performance according to the results ofFIG. 8;

FIG. 10a is a schematic representation of OFDM symbols with a guardinterval matched to an expected channel delay spread produced for asingle frequency transmission network; FIG. 10b is a schematicrepresentation of OFDM symbols with different numbers of sub-carriersper OFDM symbol with a guard interval selected as a fixed fraction ofthe related OFDM symbol duration; and FIG. 10c is a schematicrepresentation of OFDM symbols with a different number of sub-carriersper payload data bearing OFDM symbol and a different number ofsub-carriers for a signalling OFDM symbol with guard interval selectedto have a duration which is matched to both the payload and thesignalling OFDM symbols;

FIG. 11a is a schematic block diagram of a receiver for detecting andrecovering signalling data from a signalling OFDM symbol according tothe present technique, FIG. 11b is a schematic block diagram of afrequency synchronisation detector which forms part of FIG. 11a , FIG.11c is a schematic block diagram of a preamble guard interval correlatorwhich forms part of FIG. 11b , FIG. 11d is an illustrative schematicblock diagram of a further example of a coarse frequency offsetsynchronisation detector which forms part of the receiver of FIG. 11a ,and FIG. 11e is an illustrative schematic block diagram of adifferential encoder which forms part of FIG. 11 d;

FIG. 12 is a schematic block diagram of one example of a preambledetection and decoding processor which forms part of the receiver shownin FIG. 11a , which detects and removes the signature sequence in thefrequency domain;

FIG. 13 is a schematic block diagram of one example of a preambledetection and decoding processor which forms part of the receiver shownin FIG. 11a , which detects and removes the signature sequence in thetime domain;

FIG. 14 is a schematic block diagram of an example of a signaturesequence remover which forms part of the preamble detection and decodingprocessor shown in FIG. 13;

FIG. 15a is a schematic block diagram of a matched filter, which ismatched to the signature sequence for which an example generator isshown in FIG. 7, and FIG. 15b is a schematic block diagram of asignature sequence remover forming part of the receiver shown in FIG.14;

FIG. 16a is a graphical representation of a signal formed at the outputof the matched filter; FIG. 16b is an expanded view of the graphicalrepresentation shown in FIG. 16b illustrating components of a channelimpulse response;

FIG. 17 is a schematic block diagram illustrating a circuit fordetecting a coarse frequency offset in the receiver of FIG. 11 a;

FIG. 18 is a graphical plot of the correlation output of the circuitshown in FIG. 17 for a frequency offset of −88/Tu;

FIG. 19 provides a graphical plot of bit error rate with respect tosignal to noise ratio for different code rates with and without asignature sequence added to the signalling OFDM symbol for rate one halfand rate one quarter codes;

FIGS. 20a and 20b provide graphical plots of bit error rate againstsignal to noise ratio for a 0 dB echo channel with two paths asillustrated in FIG. 20c respectively with ideal and actual channelestimation.

DESCRIPTION OF EXAMPLE EMBODIMENTS

Embodiments of the present disclosure can be arranged to form atransmission network for transmitting signals representing dataincluding video data and audio data so that the transmission networkcan, for example, form a broadcast network for transmitting televisionsignals to television receiving devices. In some examples the devicesfor receiving the audio/video of the television signals may be mobiledevices in which the television signals are received while on the move.In other examples the audio/video data may be received by conventionaltelevision receivers which may be stationary and may be connected to afixed antenna or antennas.

Television receivers may or may not include an integrated display fortelevision images and may be recorder devices including multiple tunersand demodulators. The antenna(s) may be inbuilt to television receiverdevices. The connected or inbuilt antenna(s) may be used to facilitatereception of different signals as well as television signals.Embodiments of the present disclosure are therefore configured tofacilitate the reception of audio/video data representing televisionprograms to different types of devices in different environments.

As will be appreciated, receiving television signals with a mobiledevice while on the move may be more difficult because radio receptionconditions will be considerably different to those of a conventionaltelevision receiver whose input comes from a fixed antenna.

An example illustration of a television broadcast system is shown inFIG. 1. In FIG. 1 broadcast television base stations 1 are shown to beconnected to a broadcast transmitter 2. The broadcast transmitter 2transmits signals from base stations 1 within a coverage area providedby the broadcast network. The television broadcast network shown in FIG.1 operates as a so called single frequency network in which each of thetelevision broadcast base stations 1 transmit the radio signalsconveying audio/video data contemporaneously so that these can bereceived by television receivers 4 as well as mobile devices 6 within acoverage area provided by the broadcast network. For the example shownin FIG. 1 the signals transmitted by the broadcast base stations 1 aretransmitted using Orthogonal Frequency Division Multiplexing (OFDM)which can provide an arrangement for transmitting the same signals fromeach of the broadcast stations 2 which can be combined by a televisionreceiver even if these signals are transmitted from different basestations 1. Provided a spacing of the broadcast base stations 1 is suchthat the propagation time between the signals transmitted by differentbroadcast base stations 1 is less than or does not substantially exceeda guard interval that precedes the transmission of each of the OFDMsymbols then a receiver device 4, 6 can receive the OFDM symbols andrecover data from the OFDM symbols in a way which combines the signalstransmitted from the different broadcast base stations 1. Examples ofstandards for broadcast networks that employ OFDM in this way includeDVB-T, DVB-T2 and ISDB-T.

An example block diagram of a transmitter forming part of the televisionbroadcast base stations 1 for transmitting data from audio/video sourcesis shown in FIG. 2. In FIG. 2 audio/video sources 20 generate theaudio/video data representing television programmes. The audio/videodata is encoded using forward error correction encoding by anencoding/interleaver block 22 which generates forward error correctionencoded data which is then fed to a modulation unit 24 which maps theencoded data onto modulation symbols which are used to modulate OFDMsymbols. Depicted on a separate lower arm, signalling data providingphysical layer signalling for indicating for example the format ofcoding and modulation of the audio/video data is generated by a physicallayer signalling unit 30 and after being encoded by an encoding unit 32the physical layer signalling data is then modulated by a modulationunit 24 as with the audio/video data.

A frame builder 26 is arranged to form the data to be transmitted withthe physical layer data into a frame for transmission. The frameincludes a time divided section having a preamble in which the physicallayer signalling is transmitted and one or more data transmissionsections which transmit the audio/video data generated by theaudio/video sources 20. A symbol interleaver 34 may interleave the datawhich is formed into symbols for transmission before being modulated byan OFDM symbol builder 36 and an OFDM modulator 38. The OFDM symbolbuilder 36 receives pilot signals which are generated by a pilot andembedded data generator 40 and fed to the OFDM symbol builder 36 fortransmission. An output of the OFDM modulator 38 is passed to a guardinsertion unit 42 which inserts a guard interval and the resultingsignal is fed to a digital to analogue convertor 44 and then to an RFfront end 46 before being transmitted by an antenna 48.

As with a conventional arrangement OFDM is arranged to generate symbolsin the frequency domain in which data symbols to be transmitted aremapped onto sub carriers which are then converted into the time domainusing an inverse Fourier Transform. Thus the data to be transmitted isformed in the frequency domain and transmitted in the time domain. Asshown in FIG. 3 each time domain symbol is generated with a useful partof duration Tu and a guard interval of duration Tg. The guard intervalis generated by copying a part of the useful part of the symbol in thetime domain. By correlating the useful part of the burst with the guardinterval, a receiver can be arranged to detect the useful part of theOFDM symbol Tu, from which data can then be recovered from an OFDMsymbol by triggering a Fast Fourier Transform to convert the time domainsymbol samples into the frequency domain. Such a receiver is shown inFIG. 4.

In FIG. 4 a receiver antenna 50 is arranged to detect an RF signal whichis passed via a tuner 52 and converted into a digital signal using ananalogue to digital converter 54 before the guard interval is removed bya guard interval removal unit 56. After detecting the optimum positionfor performing a fast Fourier Transform (FFT) to convert the time domainsamples into the frequency domain, an FFT unit 58 transforms the timedomain samples to form the frequency domain samples which are fed to achannel estimation and correction unit 60. The channel estimation andcorrection unit 60 then estimates the transmission channel for exampleby using pilot sub-carriers which have been embedded into the OFDMsymbols. After excluding the pilot sub-carriers, all the data-bearingsub-carriers are fed to a symbol de-interleaver 64 which de-interleavesthe sub-carrier symbols. A de-mapper unit 62 then extracts the data bitsfrom the sub-carriers of the OFDM symbol. The data bits are fed to a bitde-interleaver 66, which performs the de-interleaving so that the errorcorrection decoder can correct errors in accordance with a conventionaloperation.

Framing Structure

FIG. 5 shows a schematic of the framing structure according to anexample embodiment of the present technique. FIG. 5 illustratesdifferent physical layer frames, some targeted for mobile receptionwhilst others are targeted for fixed roof-top antenna reception. Thesystem can be expanded in future to incorporate new types of frames, forthe current system, these potential new types of frames are simply knownas future extension frames (FEFs).

One requirement for fixed reception frames is an improved spectralefficiency which may be assured by such features as adopting a higherorder modulation, for example 256QAM, and higher code rates, for examplegreater than half rate, because of relatively benign channel conditions,and a high number of sub-carriers per OFDM symbol (FFT size) such as32K. This reduces the capacity loss due to the guard interval fraction.However, a higher number of sub-carriers can make such OFDM symbolsunsuitable for mobile reception because of lower tolerance to highDoppler frequency of the received signal. On the other hand, the mainrequirement for mobile reception frames could be robustness in order toensure a high rate of service availability. This can be improved byadopting such features as a low order modulation for example QPSK orBPSK, low code rates, a low number of sub-carriers per OFDM symbol (FFTsize) and a high density scattered pilot pattern etc. A low number ofsub-carriers for OFDM symbols can be advantageous for mobile receptionbecause a lower number of sub-carriers can provide a wider sub-carrierspacing and so more resilience to high Doppler frequency. Furthermore ahigh density pilot pattern eases channel estimation in the presence ofDoppler.

The framing structure shown in FIG. 5 is therefore characterised byframes which may each include payload data modulated and encoded usingdifferent parameters. This may include for example using different OFDMsymbol types having different number of sub-carriers per symbol, whichmay be modulated using different modulation schemes, because differentframes may be provided for different types of receiver. However eachframe may include at least one OFDM symbol carrying signalling data,which may have been modulated differently to the one or more OFDMsymbols carrying the payload data. Furthermore the signalling OFDMsymbol may be a different type to the OFDM symbol(s) carrying thepayload data. The signalling data is required to be recovered so thatthe payload data may be de-modulated and decoded.

What Characteristics for the Preamble?

To delimit frame boundaries, a frame preamble symbol such as the P1symbol in DVB-T2 is required. The preamble symbol would carry signallingthat describes how the following frame is built. It is expected that allof the types of receiver mentioned above whether mobile or with a fixedantenna should be able to detect and decode the preamble in order todetermine whether or not they should decode the payload in the followingframe. Desirable characteristics for such a preamble include:

-   -   1. High Capacity of Signalling; The preamble should have a high        signalling capacity—unlike the P1 preamble in DVB-T2 with        capacity of 7 signalling bits, a preamble more like in DVB-C2        with 100s of signalling bits is desirable. This suggests that        the preamble symbol should be an OFDM symbol with enough        sub-carriers to carry all the signalling information.    -   2. Common Macro-structure; All frame preambles should have a        common pre-defined macro-structure that is understood by all        receiver types. This means that the preamble symbol should have        for example a constant duration, constant number of sub-carriers        and guard interval for all frame types. This forces a constraint        that the guard interval must be similar in duration to the        longest guard interval that may be used in fixed antenna        reception, otherwise when the network uses this longest guard        interval, the preamble symbol will suffer from excessive        inter-symbol interference (ISI) and perhaps suffer decoding        failure.    -   3. Low complexity detection and decoding: The preamble symbol        detection and decoding complexity should be low enough to easily        implement in battery powered mobile receivers so as to make        efficient use of limited stored power. This constrains the        maximum FFT size and maximum FEC block length.    -   4. The preamble should be easily detected in the time domain; in        DVB-C2, all OFDM symbols within the frame structure use 4K        subcarrier spacing. This means that the receiver can start with        OFDM symbol time synchronisation followed by frequency domain        frame synchronisation (preamble detection). In an embodiment of        the present disclosure frames can be arranged such that OFDM        symbols in different physical layer frames may have difference        subcarrier spacing. Frequency domain frame synchronisation        (preamble detection) is thus not readily possible. The preamble        symbol must therefore be detected in the time domain. It is only        after the preamble is decoded and its signalling payload        interpreted that frequency domain processing of the frame can        proceed because only then would the receiver have knowledge of        the OFDM parameters (number of sub-carriers, guard interval) etc        of the data payload bearing OFDM symbols in the body of the        frame.    -   5. Robustness; The preamble should be detectable and decodeable        by all receiver types under all channel conditions where such        receivers are expected to work. This means that the preamble        should be robust to both high levels of noise, low signal to        noise ratios and high levels of Doppler shift as experienced        during reception on the move. Robustness to high levels of noise        constrains the maximum transmission parameters for coding and        modulation (MODCOD) that can be used for carrying the signalling        payload of the preamble whilst robustness to Doppler constrains        the minimum sub-carrier spacing of the preamble OFDM symbol. The        preamble OFDM symbol must use a sub-carrier spacing that is        large enough to be reasonably resilient to a high Doppler        spread. Furthermore, the preamble OFDM symbol should also allow        decoding in the presence of frequency shift, common phase error,        maximum expected multipath delay spreads etc.

As explained above the preamble OFDM symbol conveys signalling datawhilst the OFDM symbols within the body of the transmission frame conveypayload data as shown in FIG. 5. Each transmission frame shown in FIG. 5has particular characteristics. A data bearing frame 100 carries a frameof data, which may use a higher operating mode providing a higher numberof sub-carriers per OFDM symbol, for example, approximately 32 thousandsub-carriers (32 k mode) thereby providing a relatively high spectralefficiency, but requiring a relatively high signal to noise ratio toachieve an acceptable data integrity in the form of the bit error rate.The higher order operating mode would therefore be most suitable tocommunicate to stationary television receivers which have sensitivedetection capabilities including well positioned fixed antenna forrecovering audio/video data from the 32 k OFDM symbols. In contrast, theframe structure also includes a second frame 102 which is generated tobe received by mobile television receivers in a more hostile radiocommunications environment. The frame 102 may therefore be arranged toform payload bearing OFDM symbols with a lower order modulation schemesuch as BPSK or QPSK and a small or lower number of sub-carriers perOFDM symbol (FFT size) such as 4K or 8K to improve the likelihood that amobile receiver may be able to receive and recover the audio/video datain a relatively hostile environment. In both the first frame 100 and thesecond frame 102 a preamble symbol 104,106 is provided which providessignalling parameters for detecting the audio/video data transmitted inthe payload part of the transmission frame 100, 102. Similarly, apreamble symbol 108, 110 is provided for a future extension frame 112.

Design of New Preamble Symbol

Some example embodiments can provide an arrangement for forming apreamble symbol for use for example with the transmission frames shownin FIG. 5 in which there is an improved likelihood of detecting thepreamble symbol particularly in harsh radio environments. Furthermore,the framing structure shown in FIG. 5 can be devised such that thenumber of sub-carriers of the payload bearing OFDM symbols is differentfrom frame to frame and furthermore, these sub-carriers may usedifferent modulation schemes. Thus the OFDM symbols which carry thepayload data may be of a different type to the OFDM symbols carrying thesignalling data. An example block diagram of a part of the transmittershown in FIG. 2 for transmitting the signalling data is shown in FIG. 6.

In FIG. 6 the signalling data is first fed to a scrambling unit 200which scrambles the signalling data which is then fed to a forward errorcorrection (FEC) and modulator unit 202 which encodes the signallingdata with a forward error correcting code and then interleaves it beforemapping the encoded data onto a low order modulation constellation suchas BPSK, DBPSK, π/4-BPSK and QPSK. A pilot insertion unit 204 theninserts pilots between modulation symbols to form one of the OFDMsymbols of the preamble 104, 106, 108, 110. The OFDM symbol forming thepreamble is then scaled by a scaling unit 206 in accordance with apredetermined factor (1−G). The scaling unit 206 adapts the transmissionpower of the preamble with respect to a signature sequence which iscombined with the OFDM symbol of the preamble before transmission sothat the total transmission power of the preamble remains the same as itwould have been without the signature sequence.

According to the present the technique a signature sequence generator208 is configured to generate a signature sequence which is fed to asecond scaling unit 210 which scales the signature sequence by apredetermined factor G before the scaled signature sequence is combinedwith the OFDM symbol of the preamble by a combining units 212. Thus thesignature sequence W(k) is combined with the OFDM symbol in thefrequency domain so that each of the coefficients of the signaturesequence is added to one of the subcarrier signals of the OFDM symbol.The combined preamble OFDM symbol and signature sequence are thentransformed from the frequency domain to the time domain by an inverseFourier transform processor (IFFT) 214 before a guard interval insertionunit inserts a time domain guard interval. At an output of the guardinsertion unit 216 the preamble symbol is formed on output channel 218.

As can be seen for the example shown in FIG. 6 the signature sequence iscombined with the OFDM symbol carrying signalling data in the frequencydomain so that a spectrum of the preamble symbol after combining remainswithin a spectral mask for the transmission channel. As will beappreciated for some examples the signature sequence may be combinedwith the OFDM symbol in the time domain. However other bandwidthlimiting processes must then be introduced after the combination of thesignature sequence with the preamble OFDM symbol in the time domainwhich may affect the correlation properties of the signature sequence atthe receiver.

In the example illustration in FIG. 6, the scrambling of the signallingdata by the scrambling unit 200 ensures that the peak-to-average powerratio (PAPR) of the preamble symbol will not be excessive due to manysimilarly modulated OFDM sub-carriers. The scrambled signalling bits arethen forward error correction encoded by the FEC and BPSK unit 202 witha code such as 4K LDPC code at a low code rate (1/4 or 1/5) prior tomapping with a low order constellation such as BPSK, π/4-BPSK, DBPSK andQPSK within the unit 202. The pilots inserted at this stage by the pilotinsertion unit 204 are not for channel estimation, but for frequencyoffset estimation as will be explained shortly. At this stage, a complexpreamble signature sequence also composed of the same number of complexsamples as the useful sub-carriers as the OFDM symbol is added to thesamples of the signalling OFDM symbol by the combiner 212. Whengenerated, each preamble signature sequence sample is a point on theunit circle but before addition to the preamble OFDM symbol, each sampleis scaled by a predetermined factor G, by a scaler 210 and thecorresponding OFDM symbol sample is scaled by (1−G) by a scaler 206 sothat the power of the composite preamble symbol should be the same asthe power of the signalling OFDM symbol at point A in FIG. 6.

The IFFT 214 then forms the OFDM symbol in the time domain, which isthen followed by the insertion of the guard interval by the guardinsertion unit 216 which prepends the Ng samples of the preamble OFDMsymbol at the start of the preamble OFDM symbol—also known as the cyclicprefix of the preamble OFDM symbol. After guard interval insertion, apreamble OFDM time domain symbol of duration Ts=Tu+Tg made up ofNs=Nu+Ng complex samples where Tu is the useful symbol period with Nusamples and Tg is the guard interval duration with Ng samples is formed.

The Signature Sequence Generator

As explained above, the preamble symbol generator of FIG. 6 generates asignature sequence which is combined with the signalling OFDM symbol(first OFDM symbol), which forms the preamble symbol of the frame, inorder to allow a receiver to detect the preamble at lower signal tonoise ratios compared to signal to noise ratios which are required todetect and recover data from OFDM symbols carrying the payload data. Thesignature sequence generated by the signature sequence generator 208 canbe formed using two pseudo random sequence generators, one for thein-phase and other for the quadrature phase component.

In one example the signature sequence is a constant amplitude zeroautocorrelation (CAZAC) or Zadoff and Chu sequence. In other examplesthe signature sequence is formed from a pair of Gold code sequenceschosen because of their good auto-correlation properties, or otherexample signature sequence could be used such as from a pair ofM-sequences.

One example of the signature sequence generator 208 shown in FIG. 6 isshown in more detail in FIG. 7. FIG. 7 is arranged to generate a complexsignature sequence which is added to the complex samples of thesignalling OFDM symbol by the combiner 212 shown in FIG. 6.

In FIG. 7 two linear feedback shift registers are used in each case togenerate a pair of pseudo random bit sequences for the in-phase 300.1and 300.2 and quadrature 302.1 and 302.2 components. In each case, thepseudo-random bit sequence pair is combined using exclusive-OR circuits310, 312 to produce the Gold sequences for the in-phase (300.1 and300.2) and quadrature (302.1 and 302.2) part of the signature sequence,respectively. A binary to bipolar mapper unit 314, 316 then formsrespectively a sample for the in-phase 318 and quadrature (imaginary)320 components of the signature sequence. Effectively, the arrangementshown in FIG. 7 generates Gold codes formed by XORing two m-sequences.The m-sequences are generated by the linear feedback shift registers300, 302. A table 1 below shows the generator polynomials for the linearfeedback shift registers according to the example shown in FIG. 7:

TABLE 1 Generator polynomials for complex signature sequence. SequenceName Generator polynomial R_seq1 x¹³ + x¹¹ + x + 1 R_seq2 x¹³ + x⁹ +x⁵ + 1 I_seq1 x¹³ + x¹⁰ + x⁵ + 1 I_seq2 x¹³ + x¹¹ + x¹⁰ + 1Determining an Optimum Value for the Scaling Factor G

As shown in FIG. 6, the scaler 210 multiplies the signature sequence bya factor G and the scaler 206 multiplies the signalling OFDM symbol by afactor 1−G. As such, if the time domain signalling OFDM symbol signal isc(n) while the signature sequence signal is f(n), then the compositetransmitted preamble symbol s(n) is given by:s(n)=(1−G)c(n)+Gf(n)where G is the scaling applied to the signature sequence. The signaturesignal effectively adds distortion to the signalling OFDM symbol therebyincreasing the bit error rate of the signalling OFDM symbol at thereceiver. Furthermore, with a normalised power of 1, the compositesymbol in effect distributes power between the signature signal and thesignalling OFDM symbol signal. With a high value for G, the signaturesignal has more power and so frame synchronisation (detection of thepreamble) at the receiver should be achieved at a lower signal to noiseratio. However, reducing the power of the signalling OFDM symbol (inorder to increase the power of the signature signal) also means thaterror-free decoding of the signalling information itself becomes moredifficult at the receiver as the signal-to-noise of the signalling OFDMsymbol has fallen. Therefore, an optimum value for G has to be acompromise between these conflicting aims. We can further defineP=(1−G)/G which is proportional to the power ratio between thesignalling OFDM symbol and the signature signal. An appropriate valuefor G can be set by experimenting with this power ratio P.

The performance of example error correction codes which may be used forprotecting the preamble symbol can be assessed in the presence ofAdditive White Gaussian Noise, using an appropriate constellation forthe signalling information. For example a QPSK modulation scheme can beused with example error correction codes. In the present example 4K LDPChalf rate and quarter rate codes were evaluated. FIG. 8 provides agraphical illustration of the performance for communicating thesignalling data using the signalling OFDM symbol for these half andquarter rate LDPC codes and shows for each code a bit error rateperformance with respect to signal to noise ratios for an additive whiteGaussian noise channel. It can be seen that at a signal to noise ratioof −3 dB and a signal to noise ratio of 1 dB, the quarter rate and halfrate codes respectively each become error free. These values of signalto noise ratios were then increased to −2 dB and 2 dB respectively andthen the signature signal added with values of P varied until a biterror rate of zero was achieved.

As will be appreciated the error correction code which may be used toprotect the signalling data carried in the preamble symbol may havecoding rates which are different to rate one-half and rate one-quarter.In some embodiments the coding rate is less than or equal toone-quarter. In one example the coding rate is one-fifth (⅕).

FIG. 9 provides a graphical plot for code rates of one quarter and onehalf showing a bit error rate for each code rate as the factor P on thex-axis and SNR fixed to −2 dB and 2 dB respectively. As can be seen fromthese results setting P=8 dB will give a bit error rate close to zero,despite the presence of the signature sequence, which has been added tothe signalling OFDM symbol. It can also be seen experimentally, thatwith this value of the factor P, preamble detection can be achieved. Avalue of P=8 dB has, therefore, been adopted for the different half andquarter rate code rates with QPSK modulated data subcarriers of thesignalling OFDM symbol. As can be seen an optimising choice for thefactor P can be chosen from the results produced.

Determining a Suitable Guard Interval Fraction

According to example embodiments of the present technique, the samepreamble symbol will delimit physical layer frames meant for both fixedand mobile reception. In the following analysis it is assumed that abroadcast transmission system, which has both types of transmissionframes will be used. As such one of the principal factors affecting thereception of payload data bearing OFDM symbols transmitted for fixedreception is spectral efficiency. As explained above, this means the useof large numbers of sub-carriers for the OFDM symbols andcorrespondingly large FFT sizes because a smaller guard intervalfraction (GIF) can be used to get a large guard interval duration (GID).A large GID can allow a broadcast system to have a greater separationbetween broadcast transmitters and can operate in environments with agreater delay spread. In other words the broadcast transmission systemis configured with a wider spacing between transmitters forming a singlefrequency network (SFN).

FIG. 10 illustrates how the selection of the guard intervals can beaffected when different operating modes providing different numbers ofsub-carriers per OFDM symbol (different FFT sizes) are used fordifferent frames in the same transmission. In contrast to the diagramshown in FIG. 5, the diagram shown in FIG. 10 is in the time domain.Three sets of OFDM symbols are shown in the time domain illustrative ofwhat may happen at the point where one frame ends and another starts ina single transmission. In FIG. 10a the duration of the last OFDM symbol402 of the ending frame is the same as that of the first OFDM symbol 404of the starting frame. The unshaded area 405 between the two OFDMsymbols 402 and 404 represents the guard interval that precedes symbol404. In FIG. 10b an example of a preamble symbol shown as the light greyarea 406 is inserted to delimit the two frames. As can be seen, thisexample preamble symbol 406 has a shorter duration than the data bearingsymbols 402 and 404 as a consequence of having a different number ofsub-carriers per OFDM symbol. Accordingly, if the GIF for the preamblesymbol is the same as for the data symbols, the guard interval durationfor the preamble symbol will not be as long as that of the data bearingsymbols. Accordingly, if the delay spread of the channel is as long asthe guard interval of the data bearing OFDM symbol 402, then thepreamble symbol 406 will suffer inter-symbol interference from the lastsymbol 402 of the previous frame. Examples shown in FIG. 10c can providean arrangement in which the guard interval fraction for the preamblesymbol is selected to the effect that the guard interval duration of thepreamble symbol 406 matches or may be longer than the guard intervalduration of the last data bearing symbol 402 of the previous frame.

According to some example embodiments the largest number of sub-carriersper symbol is substantially thirty two thousand (32K). With a 32K FFTsize in DVB-T2 for example, the largest GIF is 19/128. For 6 MHz channelraster, this represents a GID of about 709.33 us. When this GID is usedfor the frame carrying OFDM symbols targeted for fixed receivers, thepreamble OFDM symbol GID should at least be of a similar value,otherwise, the preamble symbol will suffer inter-symbol-interferencefrom the last symbol of a previous fixed reception frame.

In a 6 MHz channel raster system in which for example DVB-T2 istransmitted, an OFDM symbol having substantially four thousandsub-carriers (4K) OFDM symbol has a duration of only 2*224*8/6=597.33us. Therefore even with a GIF=1, it is not possible to get a GID of709.33 us with a 4K OFDM symbol. A table below lists possible operatingmodes that are receivable in medium to high Doppler frequencies (for themobile environment) and some possible guard intervals.

TABLE 2 Mobile FFT modes and their possible guard intervals FFT Size Tuin 6 MHz (us) GIF GID (us) Ts (us) 4K 597.33 1 597.33 1194.667 ¼ 298.671493.338 8K 1194.67 ½ 597.33 1792.005 19/32 709.33 1904.000 ¾ 896.002090.638

From the above table it can be seen that only an 8K operating mode forthe preamble OFDM symbol has GIF<1 which matches or exceeds the maximumGID for a 32K maximum number of sub-carriers of the OFDM symbol. Inconclusion therefore, embodiments of the present technique can provide anumber of sub-carriers for the signalling or preamble OFDM symbol of8192 sub-carriers, which corresponds to an 8K FFT size, for which theGIF will be about 19/32. This means that the total signalling OFDMsymbol will have a duration of Ts 1904 us. Furthermore an 8K operatingmode will have a sub-carrier spacing, which provides a mobile receiverwith a reasonable chance of detecting and recovering the signalling datafrom the preamble OFDM symbol in medium to high Doppler frequencies. Itcan be understood that in embodiments of this disclosure, the GIF of thepreamble symbol has to be chosen to have the same GID that is the sameor longer than the longest GID of the maximum FFT size available in thesystem.

Channel Estimation Considerations

As known in OFDM transmission systems such as DVB-C2, frequency domainpreamble pilots may be inserted into a preamble symbol at regularintervals for use in channel estimation and equalisation of the preamblesymbol. A density of such pilots Dx, which is the spacing in thefrequency is dependent on the maximum delay spread that can be expectedon the channel. As explained above, with a single frequency transmissionnetwork, it can be advantageous to use a larger GID. For such singlefrequency networks, a channel impulse response can have a duration whichis equal to the GID. Thus, the delay spread of the channel for preambleequalisation may be as much as the GID. When using preamble pilotsspaced by Dx subcarriers, pilot-aided channel estimation is possible fordelay spreads as long as Tu/Dx. This means that Dx must be set suchthat:T _(u) /D _(x) ≧T _(g)

Since for an 8K preamble in a 6 MHz channel, Tu=1194.67 us,

$D_{x} \leq \left\lceil \frac{T_{u}}{T_{g}} \right\rceil$

Substituting Tu=1194.67 and Tg=709.33, D_(x)≦2. This means that morethan one in every two sub-carriers of the signalling OFDM symbol wouldbecome a pilot sub-carrier. This would have the effect of cutting thecapacity of the signalling OFDM symbol by more than half. As such, thisconclusion suggests that an alternative technique should be adopted toestimate the channel impulse response rather than using frequency domainpilots.

Frequency Offset Considerations

At first detection, the signalling or preamble OFDM symbol may have tobe decoded in the presence of any tuning frequency offsets introduced bytuner 52. This means that either the signalling data should be modulatedunto the preamble OFDM symbol in a manner that reduces the effects ofany frequency offsets or resources are inserted into the preamble symbolto allow the frequency offset to be estimated and then removed prior topreamble decoding. In one example the transmission frame may onlyinclude one preamble OFDM symbol per frame so the first option isdifficult to achieve. For the second option, additional resources can bein the form of frequency domain pilot sub-carriers, which are insertedinto the OFDM so that these can be used to estimate the frequency offsetand common phase error. The frequency offsets are then removed beforethe symbol is equalised and decoded. In a similar vein to the insertionof pilots into the data payload bearing OFDM symbols, embodiments of thepresent technique can be arranged to provide within the signalling(preamble) OFDM symbol pilot sub-carriers, which can allow for theestimation of frequency offsets that are larger than the preamblesub-carrier spacing. These pilots are not spaced regularly in thefrequency dimension to avoid instances when multipath propagation mayresult in regular nulls of the pilots across the full preamble OFDMsymbol. Accordingly, 180 pilot sub-carriers can be provided across the8K symbol with the positions defined apriori. The sub-FFT bin frequencyoffset is estimated via the detection of the preamble OFDM symbolitself. Accordingly embodiments of the present technique can provide apreamble OFDM symbol in which the number of sub-carriers carrying pilotsymbols is less than the number which would be required to estimate achannel impulse response through which the preamble OFDM symbol istransmitted, but sufficient to estimate a coarse frequency offset of thetransmitted OFDM symbol.

Frequency Offset Detection at the Receiver

As explained above the preamble is formed by combining an OFDM symbolcarrying signalling data with a signature sequence. In order to decodethe signalling data, the receiver has to first detect and capture thepreamble OFDM symbol. In one example the signature sequence may bedetected using a match filter which has impulse response which ismatched to the conjugate of the complex samples of the known signaturesequence. However any frequency offset in the received signal have aneffect of modulating the output of the matched filter and preventingaccurate detection of the signature sequence using a match filter. Anexample receiver for detecting the preamble and recovering thesignalling information provided by the preamble in the presence of afrequency offset is shown in FIG. 11a . In FIG. 11a , a signal receivedfrom an antenna is converted to a baseband signal, using a conventionalarrangement as shown in FIG. 4 and fed from an input 420 respectively toa complex number multiplier 422 and a frequency synchroniser 424. Thefrequency synchroniser 424 serves to detect the frequency offset in thereceived signal r(x) and feed a measure of the offset in respect of anumber of subcarriers to an oscillator 426. The oscillator 426 generatesa complex frequency signal which is fed to a second input of themultiplier 422 which serves to introduce a reverse of the offset intothe received signal r(x). Thus the multiplier 422 multiplies thereceived signal r(x) with the output from the oscillator 426 therebycompensating or substantially reversing the frequency offset in thereceived signal so that a preamble detection and decoding unit 430 candetect the preamble OFDM symbol and recover the signalling data conveyedby the preamble which is output on output channel 432.

FIG. 11b provides an example implementation of the frequencysynchroniser 424 which forms part of the receiver shown in FIG. 11a . InFIG. 11b the received signal is fed from the input 420 to a preambleguard interval correlator 432 which generates at a first output 434 asignal providing an indication of the start of the useful part of theOFDM symbol. A second output 436 feeds the samples of the OFDM symbol toa Fourier transform processor 438, but delayed by the number of samplesNu in the useful part. The first output 434 from the preamble guardinterval correlator 432 detects the location of the guard interval andserves to provide a trigger signal from a threshold detector 440 to theFFT 438 through a channel 442 which triggers the FFT 438 to convert theNu time domain samples of the useful part of the OFDM symbol into thefrequency domain. The output of the Fourier transform processor 438 isfed to a continuous pilot (CP) matched filter unit 444, which correlatesthe pilot signals in the received OFDM symbol with respect to replicasat the receiver which are used to set an impulse response of the CPmatched filter in the frequency domain. The matched filter 444 thereforecorrelates the regenerated pilots with the received OFDM symbol andfeeds a result of the correlation to an input to a detection thresholdunit 446. The detection threshold unit 446 detects an offset in thereceived signal in terms of the number of FFT bins on channel 448 whicheffectively provides the frequency offset which is fed to the oscillator426 for correcting the offset in the received signal.

FIG. 11e provides an example of implementation of the preamble guardinterval correlator 432 and corresponds to a conventional arrangementfor detecting the guard interval. Detection is performed by crosscorrelating the samples of the received OFDM symbol with themselvesafter a delay of Nu samples with the cross correlation outputsaccumulated over consecutive Ng sample intervals. Thus the receivedsignal is fed from an input 420 to a multiplier 450 and a delay unit 452which feeds an output to a complex conjugator 454 for multiplying by themultiplier 450 with the received signal. A delay unit 456 delays thesamples by the number of samples Ng in the guard interval and a singledelay unit 458 delays an output of an adder 460. The adder 460 receivesfrom the multiplier 450 the results of multiplying the received signalwith a conjugate of the delayed samples corresponding to the usefulsamples Nu which is then fed to the adder 460. Together adder 460, delayblocks 456 and 458 implement a moving average filter of order Ng whoseeffect is to accumulate successive outputs of the cross-correlator overNg samples. Thus at a point 434 there is provided an indication of thedetection of the useful part of the OFDM symbol by detecting the guardinterval period. The output 436 provides the delayed received signalsamples which are fed to the FFT in order to trigger the Fouriertransform after the guard interval has been detected by the first output434.

FIG. 11d provides another example of implementation of the frequencysynchroniser 424 and corresponds to a first detection of the preamblesymbol by use of a signature sequence matched filter 462. Firstlyhowever, the differential encoder block 461 is used to alter thereceived signal so as to reduce the modulation of the matched filteroutput by any frequency offset present in the received signal. Thedifferential encoder 461 is applied both to the received signal and thetime domain signature sequence which is generated by inverse Fouriertransform 506 of the output of the frequency domain signature sequencegenerator 504. The signature sequence matched filter 462 to be describedlater in FIG. 15a is a finite impulse response filter whose taps are setto the coefficients of the differential encoded time domain signaturesequence. The circuit shown in FIG. 11d therefore forms an example ofthe frequency synchroniser 424 in which the signature sequence generator504 re-generates the signature sequence, the inverse Fourier transformer506 transforms the signature sequence into the time domain, and thedifferential encoder 461 compares differentially successive samples ofthe received signal to reduce a modulating effect of the frequencyoffset in the radio signal, and correspondingly compares differentiallysuccessive samples of the time domain version of the signature sequence.As already explained, the matched filter 462 has an impulse responsecorresponding to the differentially encoded signature sequence andreceives the received signal from the differential encoder 461 andfilters the differentially encoded received signal to generate at anoutput an estimate of the coarse frequency offset.

Corresponding to output channel 434 in FIG. 11b , output channel 463 inFIG. 11d produces a signal which is fed to the threshold block 440 togenerate a trigger for the FFT 438; whilst output channel 436 in FIG.11b corresponds to output channel 464 in FIG. 11d . This channel conveysthe preamble OFDM symbol samples to the FFT block 438 which at the rightmoment is triggered by through channel 442 by the threshold block 440.FIG. 11e provides an example of the differential encoding block 461. Thereceived samples r(n) enter a unit delay element 465 and also aconjugation block 466. The delay element 465 delays each sample for onesample period while the conjugation element 466 changes each inputsample to its conjugate at its output whose effect is to convert aninput [r_(i)(n)+jr_(q)(n)] into an output [r_(i)(n)−jr_(q)(n)]. Thisconjugated sample is then subtracted from the output of delay element465 by the adder 467. For an input signal [r_(i)(n)+jr_(q)(n)] andoutput [y_(i)(n)+jy_(q)(n)] n=0, 1, 2 . . . , the differential encoder461 acts to implement the equation:[y _(i)(n)+jy _(q)(n)]=[r _(i)(n−1)−r _(i)(n)]+j[r _(q)(n−1)+r _(q)(n)]

Accordingly before preamble detection and decoding is performed by thepreamble detection and decoding unit 430 the frequency offset in thereceived signal is estimated and corrected by the arrangements shown inFIGS. 11a and 11 b and 11 c; or 11 d and 11 e.

Preamble Detection and Decoding at the Receiver

As explained above for the example of the receiver shown in FIG. 11a , apreamble detector and decoder 430 is configured to detect the preamblesymbol and to recover the signalling data from the preamble symbol. Tothis end, the preamble detector and decoder 430 detects the preamble bydetecting the signature sequence and then removes the signature sequencebefore recovering the signalling data from the preamble. Exampleembodiments of the preamble detector and decoder 430 are illustrated inFIGS. 12, 13 and 14.

Embodiments of the present technique can provide a receiver whichdetects the signature sequence and removes the signature sequence in thefrequency domain or in the time domain. FIG. 12 provides a first examplein which the signature sequence is removed in the frequency domain.Referring to the example receiver shown in FIG. 11a , the received baseband signal is fed from a receive channel 428 to a matched filter 502and a demodulator 550. The match filter 502 receives the signaturesequence in the time domain after a signature sequence generator 504,which is the same as the signature sequence generator 212 at thetransmitter, re-generates a copy of the signature sequence. The matchedfilter 502 is configured to have an impulse response which is matched tothe time domain signature sequence. As such, it correlates the timedomain signature sequence with the received signal fed from the receivechannel 428 and the correlation output result can be used to detect thepresence of the preamble OFDM symbol when an output of the correlationprocess exceeds a predetermined threshold. Furthermore, as a result ofthe presence of the signature sequence in the preamble OFDM symbol, animpulse response of the channel through which the received signal haspassed can also be estimated from the correlation output of the matchedfilter by a channel impulse response estimator 508. The receiver cantherefore include an arrangement for estimating the channel impulseresponse using the signature sequence without recourse to thetraditional scattered pilots.

Having detected the presence of the signature sequence and estimated thechannel impulse response, the effect of the channel impulse response canbe removed from the received signal within the demodulator 550.Accordingly a Fast Fourier Transformer 518 transforms the channelimpulse response estimate into the frequency domain channel transferfunction and feeds the channel transfer function to an equaliser 516within the demodulator 550.

In the receiver shown in FIG. 12 the demodulator 550 is arranged torecover the signalling data in a base band form encoded with an errorcorrection code. The demodulator 550 therefore recovers the signallingdata from the signalling (preamble) OFDM symbol, which is then decodedusing a forward error correction decoder 520 before being descrambled bya descrambling unit 522 which corresponds to the scrambling unit 200shown in FIG. 6 but performs a reverse of the scrambling.

The demodulator 550 includes a guard interval remover 512, which removesthe guard interval from the signalling OFDM symbols, and an FFT unit514, which converts the time domain samples into the frequency domain.The equaliser 516 removes the effects of the channel impulse response,which has been converted into the frequency domain to form a channeltransfer function by the FFT unit 518 as already explained above. In thefrequency domain the equaliser 516 divides each signalling data carryingOFDM sub-carrier by its corresponding channel transfer coefficient toremove, as far as possible, the effect of the transmission channel fromthe modulation symbols.

A signature sequence remover is formed by an adder unit 519 whichreceives the signature sequence in the frequency domain generated by thesignature sequence generator 504 after this has been scaled by thescaling factor G, as explained above by a scaling unit 521. Thus thesignature sequence remover 519 receives at a first input the equalisedpreamble OFDM symbol and on a second input a scaled signature sequencein the frequency domain and subtracts one from the other to form at theoutput estimates of the modulation symbols which were carried by thedata bearing subcarriers of the preamble OFDM symbol.

The modulation symbols representing the error correction encodedpreamble signalling data are then demodulated and error correctiondecoded by the demodulator and FEC decoder 520 to form at an output thescrambled bits of the L1 signalling data which are then descrambled bythe descrambling unit 522 to form as an output 524 the L1 signallingdata bits.

A further example of the preamble detector and decoder 430 whichoperates in the time domain to remove the signature sequence is showingin FIGS. 13 and 14. FIG. 13 provides an example of the preamble detectorand decoder 430 which corresponds to the example shown in FIG. 12 and soonly differences with respect to the operation of the example shown inFIG. 13 will be explained. In FIG. 13 as with the example in FIG. 12 thebaseband received signal is fed to a signature sequence matched filter502 and to a demodulator 550. As with the example shown in FIG. 12, thesignature sequence matched filter cross-correlates the received signalwith an impulse response which is matched to the time domain signaturesequence. The signature sequence is received in the time domain form byregenerating the signature sequence in the frequency domain using thesignature sequence generator 504 and transforming the signature sequenceinto the time domain using an inverse Fourier transform processor 506.As with the example shown in FIG. 12 a channel impulse responseestimator 508 detects the channel impulse response from the output ofthe signature sequence matched filter 502 and forms this into thefrequency domain channel transfer function using an FFT unit 518 to feedthe frequency domain channel estimate to an equaliser 516 within thedemodulator 550.

So far the operation of the example shown in FIG. 13 corresponds to thatshown in FIG. 12. As shown in FIG. 13 the demodulator 550 includes thesignature sequence remover 559 at before the guard remover 512. The timedomain signature sequence which is fed from the inverse Fouriertransform unit 560 is scaled by the scaling unit 521 by thepredetermined factor G. The scaled time domain signature sequence isthen fed to the signature sequence remover 559 which removes thesignature sequence in the time domain from the received baseband signal.Thereafter the guard remover 512, the FFT unit 514 and the equaliser 516operate in a corresponding way to the elements shown in FIG. 12.

The signature sequence remover 559 shown in FIG. 13 is shown in moredetail in FIG. 14. In FIG. 14 the signature sequence remover 559comprises a guard interval inserter 561, a combiner unit 560 and an FIRfilter 562. The time domain baseband received signal is received on theinput channel 428 at one input of the combiner unit 560. A second input564 receives the scaled time domain version of the signature sequence,which is fed to the guard interval inserter 561 which prepends a cyclicprefix to the signature sequence in much the same way as the guardinterval inserter 561 42 at the transmitter. The output of the guardinterval inserter feeds the FIR filter 562 which receives on a secondinput 566 the estimate of the channel impulse response generated by thechannel impulse response extraction block 508. The FIR filter 562therefore convolves the channel impulse response estimate with thesignature sequence in the time domain which is then subtracted by thecombiner 560 from the received baseband signal to remove the effect ofthe signature sequence from the received signal. FIG. 15b shows a moredetailed example implementation of this signature sequence removal andhow the FIR filter 562 is configured.

As will be appreciated the operation of the demodulator and FEC decoder520 and the scrambler 522 perform the same functions as explain withreference to FIG. 12.

Matched Filter

As indicated above the matched filter 502 generates an output signalwhich represents a correlation of the received signal with the signaturesequence. A block diagram showing an example of the signature sequencematched filter 502 is shown in FIG. 15 a.

FIG. 15a shows a sequence of Ns delay elements 600 connected to scalingunits 602 which scale each of the samples of the data stored in thedelay storing unit 600 by a corresponding component of the signaturesequence P(n) but conjugated. The output from each of the scaling units602 is then fed to an adding unit 604 which forms an output signalrepresenting a correlation of the received signal samples r(n) with thesignature sequence at an output 606. The matched filter implements theequation:g(i)=Σ_(n=0) ^(N) ^(s) ⁻¹ P*(n)r(n+i) for i=−Ns+1,−Ns+2 . . . ,0,1,2, .. . Ns−1

When the filter taps P(i) are of form (±1±j1), the multiplier at eachtap could simply be done by add and subtract circuits for each of thein-phase and quadrature components. When the signature sequence is aCAZAC sequence, the quadrature components of P(i) are not bipolar. Thescaling units 602 can use the sign of each quadrature component insteadso as to have the form (±1±j1).

FIG. 16a and FIG. 16b provide examples of a correlation output of thematch filter for a multipath environment. In this case the channel iscomposed of three paths and the preamble is a 4K symbol with GIF of ¼for illustrative purposes only. As can be seen there is a clearcorrelation peak when the signature sequence of the received signalcoincides with the signature sequence at the receiver. The example shownin FIG. 16b shows the output of the match filter but with a moreexpanded x-axis showing an increase in resolution which is expanded fromthe correlation peak shown in FIG. 16a . For this channel, there arethree paths as tabulated in the Table below:

TABLE 3 Multipath profile of example channel Path Delay (us) [samples]Power (dB) 1 0 [0] 0 2 10 [68] −10 3  25 [171] −6Channel Impulse Response Extractor

As can be seen from FIG. 16b , both the amplitudes of the main impulsesand their relative delays coincide with the characteristics of themultipath channel profile through which this particular signalpropagated. To detect the actual channel paths, a threshold of energydetection is set to an appropriate multiple of the root mean square(RMS) level of the matched filter output within a window ±Ns of thehighest amplitude output sample. The exact multiple of the RMS is chosenexperimentally depending on the lowest signal to noise ratio under whichthe system is to work. Any sample of the matched filter output abovethis threshold is taken as a channel path, and all other samples arethen set to zero in the channel impulse estimator 508. Finally, thechannel impulse response (CIR) is normalised by dividing all its sampleswith the highest amplitude sample. In this way, the relative amplitudesand delays of each of the impulses in the channel through which thereceived signal has passed can be estimated.

Signature Sequence Remover

Having formed an estimate of the channel impulse response, a componentof the received signal corresponding to that contributed by thesignature sequence in the received signal can be generated by passingthe received signal r(i) through the signature sequence remover 559,which is configured with filter taps h_(n) to reflect the delay andamplitude profile of the channel impulse response. This can beaccomplished by suitable scaling, shifting and adding of the signaturesequence of length Ns=Nu+Ng of the preamble symbol. An example of thefilter is shown in FIG. 15 b.

As shown in FIG. 15b , the signature sequence remover 559 includes afinite impulse response (FIR) filter 562 made up of a delay linecomprised of Ns−1 delay elements 652.1, 652.2, to 652.Ns−1. The outputof these delay elements are connected to corresponding gain terms 651.1,651.2, to 651.Ns−1 each of which gain stages feed their output to theadder 653. The input 654 of the filter is connected both to the input ofdelay element 652.1 and to the input of gain term 651.0. The output 656of the FIR filter 650 is connected to the input of an adder 560 whoseother input 657 receives the received preamble signal samples r(i).During operation, the gain stages of the FIR filter are set to thenegative values of the samples of the channel impulse response derivedby the channel impulse response estimator 506. The FIR 650 generates atan output 656 a signal representing the convolution of the signaturesequence by the channel impulse response estimate, which effectivelyprovides an estimate of the effect of the channel on the signaturesequence imposed upon the signalling OFDM symbol. An adder 560 thensubtracts the output signal of the FIR 656 from the received signal froman input 657 to remove the effect of the signature sequence from thereceived signal to form an output 660. Therefore a result (of thesignature sequence transiting the channel described by the channelimpulse response) is subtracted from the received signal by thesignature sequence remover 510 with a delay matched to the point fromwhich the first significant impulse (of the output of the matchedfilter) occurred. This process can be iterated in that the matchedfilter 502 can be re-run with the results of the subtraction, thechannel impulse response re-estimated by the channel impulse responseestimator 508 and the its effect on the signature sequence beingextracted again by the signature sequence remover 559. As a result, amore accurate estimate of the effect of the signature sequence on thereceived signal can be estimated and subtracted from the receivedsignal. Channel impulse responses from all iterations can then be summedand normalised to provide an improved estimate of the channel impulseresponse from which the channel transfer function (CTF) is derived forpreamble symbol equalisation.

Frequency Offset Estimation

FIG. 17 provides a more detailed schematic block diagram of the preamblepilot matched filter 444 used for detecting a coarse frequency offset inthe received signalling OFDM symbol, which may form part of thefrequency synchroniser 424 of FIG. 11a , As explained above, the numberof pilots introduced into the signalling OFDM symbol is less than thenumber which would be required in order to estimate the channel. Thenumber of pilot symbols is therefore set to estimate a coarse frequencyoffset. The block diagram shown in FIG. 17 provides an examplerepresentation of the coarse frequency remover 513 and is shown withthree versions of the received preamble signal 701.

As shown in FIG. 17 a sequence of delay elements 700 are used to feed indiscrete samples of the signal which are then multiplied by multipliers702 with the known pilot signal values P(n) and summed by a summing unit704 to form a correlation output 706. A pulse detector or peak detector708 is the same one shown as 446 in FIG. 11b which then generates anoutput signal on channel 710 showing a peak when there is a coincidencebetween a relative offset of the received signal with the company of thepilot signals at the receiver. Shaded circles of each received signal701 show sub-carrier cells that represent preamble pilots whilst theun-shaded cells show non-pilot sub-carrier cells. All sub-carrier cellsare shifted into the transversal filter from right to left. Theparameter MaxOff is a design parameter that represents the maximum valueof the frequency offset in units of sub-carrier spacing Ω that thedesigner may expect. The output of the pulse detector is only validbetween shifts (0.5(Na+Nu)−MaxOff) and (0.5(Na+Nu)+MaxOff) where Na isthe number of sub-carriers (out of a total of Nu) used in the preambleOFDM symbol. If the shifts are numbered from −MaxOff to +MaxOff then thepulse detector output will go high for the shift that corresponds to theobserved frequency offset.

Once Ω is detected, this coarse frequency is removed by shifting thesubcarriers by −Ω i.e. in the opposite direction to the frequencyoffset. This can also be removed prior to FFT in common with the finefrequency offset which is estimated from the argument of the peakpreamble detection matched filter or guard interval correlation 432 peaksample by modulation with a suitably phased sinusoid generated by theoscillator 426 in FIG. 11a . The two frequency offsets can be used tostart off the carrier correction loop for the rest of the OFDM symbolsin the frame.

FIG. 18 shows a pilot correlation result of a frequency offset in anexample plot of the input of the pulse detector for a frequency offsetof Ω=−88 in a case where MaxOff is set to 350. The pulse detector mightuse a threshold to clip this input as a detector of the presence orabsence of a substantial pulse.

Preamble Symbol Equalisation

After signature sequence removal from the received samples and thecoarse frequency offset has been adjusted, OFDM equalisation can beginwith the FFT of the received sequence. The FFT window starts from atrigger position in the FFT unit 514 corresponding to the relative delayof the first impulse in the channel impulse response estimate. If thechannel impulse response estimate duration is longer than the preambleGID, then the trigger position is altered to ensure that it starts atthe beginning of a Ng (Ng is the number of time domain samples in theguard interval of the preamble symbol) long window under which themaximum of the energy of the channel impulse response estimate falls.The Nu point FFT produces the preamble OFDM symbol in the frequencydomain with the effect of the channel superposed. Before equalisationand decoding, any frequency offsets have to be calculated and removed bythe frequency offset remover as explained above with reference to FIGS.11a, 11b, 11c . This estimation uses correlation with the known preamblepilots to determine how far to the right or left the full symbol isshifted in frequency. Equalisation of the preamble OFDM symbol requiresa channel transfer function (CTF). This is derived by executing a Nupoint FFT on the channel impulse response estimate by the FFT unit 518.This provides a channel transfer function for all subcarriers in thepreamble OFDM symbol allowing subcarrier by subcarrier one-tapequalisation to take place. Finally, the equalised data subcarriers areextracted (pilot subcarriers discarded) and de-mapped, forward errorcorrection (FEC) decoded to provide the signalling.

Selected Results

FIG. 19 provides a graphical plot of bit error rate with respect tosignal to noise ratio for different code rates with and without theaddition of the signature sequence to the signalling OFDM symbol. Thus,two code rates are shown, rate one half and rate one quarter, each coderate including the example of the presence of the signature sequence andwithout the signature sequence. As can be seen, the results for rate onequarter show that the signalling OFDM symbol can be detected even atsignal to noise ratios of less than −2 dBs.

Two further sets of results shown in FIGS. 20a and 20b provide agraphical plot of bit error rate against signal to noise ratio in whichfor the results shown in FIG. 20a there is a 0 dB echo channel with anideal channel estimation and in FIG. 20b a multipath environment withtwo paths as illustrated in FIG. 20c . Thus for FIG. 20b in contrast tothe result shown in FIG. 20a there is a relative degradation inperformance resulting from real channel estimation. However, as can beseen, the results are comparable.

The following numbered clauses provide further example aspects andfeatures of the present disclosure:

1. A transmitter for transmitting payload data using OrthogonalFrequency Division Multiplexed (OFDM) symbols, the transmittercomprising

a frame builder configured to receive the payload data to be transmittedand to receive signalling data for use in detecting and recovering thepayload data at a receiver, and to form the payload data with thesignalling data into frames for transmission,

a modulator configured to modulate a first OFDM symbol with thesignalling data forming a part of each of the frames and to modulate oneor more second OFDM symbols with the payload data to form each of theframes,

a combiner for combining a signature sequence with the first OFDMsymbol, and

a transmission unit for transmitting the first and second OFDM symbols,wherein the first OFDM symbol carrying the signalling data is combinedbefore transmission with the signature sequence.

2. A transmitter according to clause 1, wherein the first OFDM symbol isa first type having a number of sub-carriers, which is less than orequal to the number of sub-carriers of the one or more second OFDMsymbols of a second type.

3. A transmitter according to clause 1 or 2, comprising a signaturesequence generator for generating the signature sequence, wherein thesignature sequence is configured to be detected by a receiver fordetecting the first OFDM symbol within the frame before the one or moresecond OFDM symbols.

4. A transmitter according to clause 1, 2 or 3, wherein the signaturesequence comprises a set of complex coefficients and the signaturesequence is combined with the first OFDM symbol by adding each of thecomplex coefficients to a corresponding one of the samples of the firstOFDM symbol in the time domain.

5. A transmitter according to clause 1, 2 or 3, wherein the signaturesequence comprises a set of complex coefficients and the signaturesequence is combined with the first OFDM symbol in the frequency domain.

6. A transmitter according to clause 4 or 5, wherein the set of complexcoefficients of the signature sequence is based on a sequence generatedusing at least a first pseudo-random binary sequence generatorconfigured to generate a real component of the complex coefficients, andat least a second pseudo-random binary sequence generator separatelyconfigured to generate the imaginary component of the complexcoefficients and where the pseudo-binary sequence generated in each caseis an M-sequence or Gold code or the like.

7. A transmitter according to clause 4 or 5, wherein the set of complexcoefficients of the signature sequence is formed from a constantamplitude zero autocorrelation sequence.

8. A transmitter according to any of clauses 1 to 7, wherein thesignature sequence is scaled by a first predetermined value and thefirst OFDM symbol is scaled by a second predetermined value.

9. A transmitter according to clause 8, wherein the second predeterminedfactor is equal to one minus the first predetermined factor.

10. A transmitter according to clause 8 or 9, wherein the firstpredetermined factor is set as a balance between a reduction in thelikelihood of correctly recovering the signalling data and an increasein the likelihood of detecting the signature sequence at the receiver.

11. A transmitter according to any of clauses 1 to 10, wherein thesignalling data is encoded with a first error correction code and thepayload data is encoded with at least one other error correction code,an encoding rate of the first error correction code being lower than anencoding rate of the at least one other error correction code.

12. A transmitter according to any of clauses 1 to 9, wherein thesignalling data is encoded with an error correction code, an encodingrate of the error correction code being equal to or lower than rate onequarter.

13. A method of transmitting payload data using Orthogonal FrequencyDivision Multiplexed (OFDM) symbols, the method comprising

receiving the payload data to be transmitted,

receiving signalling data for use in detecting and recovering thepayload data to be transmitted at a receiver,

forming the payload data with the signalling data into frames fortransmission, the signalling data forming a part of each of the frameswith the payload data,

modulating a first OFDM symbol with the signalling data and modulatingone or more second OFDM symbols with the payload data to form each ofthe frames,

combining the first OFDM symbol with a signature sequence, and

transmitting the frames with first and second OFDM symbols.

14. A method according to clause 13, wherein the first OFDM symbol is afirst type having a number of sub-carriers which is less than or equalto the number of sub-carriers of the one or more second OFDM symbols ofa second type.

15. A method according to clause 13 or 14, comprising generating thesignature sequence, the signature sequence being detectable by areceiver for detecting the first OFDM symbol within the frame before theone or more second OFDM symbols.

16. A method according to clause 13, 14 or 15, wherein the signaturesequence comprises a set of complex coefficients and the signaturesequence is combined with the first OFDM symbol by adding each of thecomplex coefficients with a corresponding one of the samples of thefirst OFDM symbol in the time domain.

17. A method according to clause 13, 14 or 15, wherein the signaturesequence comprises a set of complex coefficients and the signaturesequence is combined with the first OFDM symbol by adding each of thecomplex coefficients with corresponding samples of the first OFDM symbolin the frequency domain.

18. A method according to clause 17, wherein the set of complexcoefficients of the signature sequence is based on a sequence generatedusing at least a first pseudo-random binary sequence generatorconfigured to generate a real component of the complex coefficients, andat least a second pseudo-random binary sequence generator separatelyconfigured to generate the imaginary component of the complexcoefficients and where the pseudo-binary sequence generated in each caseis an M-sequence or Gold code or the like.

19. A method according to clause 16 and 17, wherein the set of complexcoefficients of the signature sequence is formed from a constantamplitude zero autocorrelation sequence.

20. A method according to any of clauses 13 to 19, comprising

scaling the signature sequence by a first predetermined value, and

scaling the first OFDM symbol by a second predetermined value.

21. A method according to clause 20, wherein the second predeterminedfactor is equal to one minus the first predetermined factor.

22. A method according to clause 20 or 21, wherein the firstpredetermined factor is set as a balance between a reduction in thelikelihood of correctly recovering the signalling data and an increasein the likelihood of detecting the signature sequence at the receiver.

23. A method according to any of clauses 13 to 22, comprising

encoding the signalling data with a first error correction code, and

encoding the payload data with at least one other error correction code,an encoding rate of the first error correction code being lower than anencoding rate of the at least one other error correction code.

24. A method according to any of clauses 13 to 22, comprising

encoding the signalling data with an error correction code, an encodingrate of the error correction code being equal to or lower than rate onequarter.

25. A receiver for detecting and recovering payload data from a receivedsignal, the receiver comprising

a detector for detecting the received signal, the received signalcomprising the payload data and signalling data for use in detecting andrecovering the payload data, the signalling data and the payload dataforming frames in the received signal, the signalling data in each framebeing carried by a first Orthogonal Frequency Division Multiplexed,OFDM, symbol, and the payload data being carried by one or more secondOFDM symbols, and the first OFDM symbol having been combined with asignature sequence,

a matched filter having an impulse response which has been matched tothe signature sequence with the effect that an output of the matchedfilter generates a signal representing a correlation of the signaturesequence with the received signal,

a synchronisation detector configured to detect the first OFDM symbolfrom the output signal of the matched filter, and

a demodulator for recovering the signalling data from the first OFDMsymbol for recovering the payload data from the second OFDM symbol.

26. A receiver according to clause 25, comprising

a channel impulse response estimator configured to detect an impulseresponse of a channel through which the received signal has passed fromthe output signal of the matched filter, wherein the demodulator isconfigured to remove the effects of the channel impulse response fromthe received signal to recover the signalling data.

27. A receiver according to clause 26, comprising a signature sequenceremover configured

to receive the estimate of the channel impulse response,

to convolve the estimate of the channel impulse response with thesignature sequence, and

to subtract the convolved channel impulse response estimate with thesignature sequence from the received signal.

28. A receiver according to any of clauses 25, 26 or 27, wherein thechannel impulse response estimator is configured to estimate the channelimpulse response by

detecting samples of the matched filter output signal which exceed apredetermined threshold,

setting samples below the predetermined threshold to zero, and

normalising the samples above the predetermined threshold with respectto the largest of the samples.

29. A receiver according to any of clauses 25 to 28, wherein thesignature sequence comprises a set of complex coefficients, the set ofcomplex coefficients of the signature sequence being based on a sequencegenerated using at least a first pseudo-random binary sequence generatorconfigured to generate a real component of the complex coefficients, andat least a second pseudo-random binary sequence generator separatelyconfigured to generate the imaginary component of the complexcoefficients and where the pseudo-binary sequence generated in each caseis an M-sequence or Gold code or the like.

30. A receiver according to clause 29, wherein the set of complexcoefficients of the signature sequence is formed from a constantamplitude zero autocorrelation sequence.

31. A receiver according to any of clauses 25 to 29, wherein thedemodulator includes

a guard interval remover configured to detect a location of a guardinterval part of the first OFDM symbol and a useful part of the firstOFDM symbol,

a forward Fourier transform configured to perform a Fourier transform onthe useful part of the first OFDM symbol, and

an equaliser configured to remove the channel response estimate from thefirst OFDM symbol in the frequency domain.

32. A receiver according to any of clauses 25 to 31, wherein thesignalling data has been encoded with a first error correction code andthe payload data is encoded with at least one other error correctioncode, an encoding rate of the first error correction code being lowerthan an encoding rate of the other error correction code, and thereceiver comprises

an error correction decoder configured to decode the first errorcorrection encoded signalling data to generate an estimate of thesignalling data.

33. A receiver according to any of clauses 25 to 32, wherein thesignalling data has been encoded with an error correction code, anencoding rate of the error correction code being lower than rate onequarter, and the receiver comprises

an error correction decoder configured to decode the error correctionencoded signalling data to generate an estimate of the signalling data.

34. A method of detecting and recovering payload data from a receivedsignal, the method comprising

detecting the received signal, the received signal comprising thepayload data and signalling data for use in detecting and recovering thepayload data, the signalling data and the payload data forming frames inthe received signal, the signalling data in each frame being carried bya first Orthogonal Frequency Division Multiplexed, OFDM, symbol, and thepayload data being carried by one or more second OFDM symbols, and thefirst OFDM symbol having been combined with a signature sequence,

filtering the received signal with a matched filter having an impulseresponse which has been matched to the signature sequence with theeffect that an output of the matched filter generates a signalrepresenting a correlation of the signature sequence with the receivedsignal,

detecting the first OFDM symbol from the output signal of the matchedfilter, and

demodulating the first OFDM symbol to recover the signalling data foruse in recovering the payload data from the second OFDM symbol.

35. A method according to clause 34, comprising

detecting an impulse response of a channel through which the receivedsignal has passed from the output of the matched filter, and

removing the effects of the channel impulse response from the receivedsignal to recover the signalling data.

36. A method according to clause 35, comprising

receiving the estimate of the channel impulse response,

convolving the estimate of the channel impulse response with thesignature sequence, and

subtracting the convolved channel impulse response estimate with thesignature sequence from the received signal, to remove an effect of thesignature sequence from the received signal.

37. A method according to any of clauses 34, 35 or 36, comprising

detecting samples of the matched filter output signal which exceed apredetermined threshold,

setting samples below the predetermined threshold to zero, and

normalising the samples above the predetermined threshold with respectto the largest of the samples.

38. A method according to any of clauses 34 to 37, wherein the signaturecomprises a set of complex coefficients, the set of complex coefficientsof the signature sequence being based on a sequence generated using atleast a first pseudo-random binary sequence generator configured togenerate a real component of the complex coefficients, and at least asecond pseudo-random binary sequence generator separately configured togenerate the imaginary component of the complex coefficients and wherethe pseudo-binary sequence generated in each case is an M-sequence orGold code or the like.

39. A method according to clause 38, wherein the set of complexcoefficients of the signature sequence is formed from a constantamplitude zero autocorrelation sequence.

40. A receiver according to any of clauses 34 to 39, wherein thedemodulating includes

detecting a location of a guard interval part of the first OFDM symboland a useful part of the first OFDM symbol,

performing a Fourier transform on the useful part of the first OFDMsymbol, and

removing the channel response estimate from the first OFDM symbol in thefrequency domain.

41. A method according to any of clauses 34 to 40, wherein thesignalling data has been encoded with a first error correction code andthe payload data is encoded with at least one other error correctioncode, an encoding rate of the first error correction code being lowerthan an encoding rate of the other error correction code, and the methodcomprises

decoding the first error correction encoded signalling data to generatean estimate of the signalling data.

42. A method according to any of clauses 34 to 41, wherein thesignalling data has been encoded with an error correction code, anencoding rate of the error correction code being lower than rate onequarter, and the method comprises

decoding the error correction encoded signalling data to generate anestimate of the signalling data.

43. A computer program providing computer executable instructions whichwhen loaded onto a computer causes the computer to perform the methodaccording to any of clauses 34 to 42 or any of clauses 13 to 24.

Various further aspects and features of the present disclosure aredefined in the appended claims. Various combinations of features may bemade of the features and method steps defined in the dependent claimsother than the specific combinations set out in the attached claimdependency. Thus the claim dependencies should not be taken as limiting.

The invention claimed is:
 1. A receiver for detecting and recoveringdata from a received signal, the receiver comprising: circuitryconfigured to receive a signal, the received signal comprising payloaddata and signalling data for use in detecting and recovering the payloaddata, the signalling data and the payload data forming frames in thereceived signal, the signalling data in each frame being carried by atleast one first Orthogonal Frequency Division Multiplexed, OFDM, symbol,and the payload data being carried by one or more second OFDM symbols,and a plurality of sub-carriers of the at least one first OFDM symbolhaving been combined with a sequence; filter the received signal with amatched filter having an impulse response which is matched to thesequence with the effect that an output of the matched filter generatesa signal representing a correlation of the sequence with the receivedsignal; detect the at least one first OFDM symbol from the output signalof the matched filter; and derive the signalling data from the detectedat least one first OFDM symbol for recovering the payload data from theone or more second OFDM symbols using the derived signalling data,wherein the sequence comprises a set of complex coefficients generatedusing at least a first pseudo-random binary sequence generatorconfigured to generate the real components of the complex coefficients,and at least a second pseudo-random binary sequence generator separatelyconfigured to generate the imaginary components of the complexcoefficients.
 2. The receiver as claimed in claim 1, wherein thecircuitry is configured to detect an impulse response of a channelthrough which the received signal has passed from the output signal ofthe matched filter, and remove the effects of the channel impulseresponse from the received signal to recover the signalling data.
 3. Thereceiver as claimed in claim 2, wherein the circuitry is configured toestimate the channel impulse response by detecting samples of thematched filter output signal which exceed a threshold, setting samplesbelow the threshold to zero, and normalising the samples above thethreshold with respect to the largest of the samples.
 4. The receiver asclaimed in claim 1, wherein the set of complex coefficients of thesequence is formed from a constant amplitude zero autocorrelationsequence.
 5. The receiver as claimed in claim 1, wherein the circuitryis configured to detect a location of a guard interval part of one ofthe at least one first OFDM symbol and a useful part of the one of theat least one first OFDM symbol, perform a Fourier transform on theuseful part of the one of the at least one first OFDM symbol, and removethe channel response estimate from the one of the at least one firstOFDM symbol in the frequency domain.
 6. The receiver as claimed in claim1, wherein the circuitry is configured to regenerate the sequence fromtwo input sequences.
 7. A method for detecting and recovering data froma received signal, the method comprising: receiving the signal, thereceived signal comprising payload data and signalling data for use indetecting and recovering the payload data, the signalling data and thepayload data forming frames in the received signal, the signalling datain each frame being carried by at least one first Orthogonal FrequencyDivision Multiplexed, OFDM, symbol, and the payload data being carriedby one or more second OFDM symbols, and a plurality of sub-carriers ofthe at least one first OFDM symbol having been combined with a sequence;filtering the received signal with a matched filter having an impulseresponse which is matched to the sequence with the effect that an outputof the matched filter generates a signal representing a correlation ofthe sequence with the received signal; detecting the at least one firstOFDM symbol from the output signal of the matched filter; and derivingthe signalling data from the detected at least one first OFDM symbol torecover the payload data from the one or more second OFDM symbols usingthe derived signalling data, wherein the sequence comprises a set ofcomplex coefficients generated using at least a first pseudo-randombinary sequence generator configured to generate the real components ofthe complex coefficients, and at least a second pseudo-random binarysequence generator separately configured to generate the imaginarycomponents of the complex coefficients.